“We thought, because we had power, we had wisdom”.
It has been mentioned several times in the earlier chapters of this book that running opamps at the slightly higher voltage of ±17 V rather than ±15 V gives an increase in headroom and dynamic range of 1.1 dB for virtually no cost and with no reliability penalty. Soundcraft ran all the opamps in their mixing consoles at ±17 V for at least two decades, and opamp failures were almost unknown. This recommendation assumes that the opamps concerned have a maximum supply voltage rating of ±18 V, which is the case for the Texas TL072, the new LM4562, and many other types.
The 5532 is (as usual) in a class of its own. Both the Texas and Fairchild versions of the NE5532 have an absolute maximum power supply voltage rating of ±22 V (though Texas also gives a “recommended supply voltage” of ±15 V), but I have never met any attempt to make use of this capability. The 5532 runs pretty warm on ±17 V when it is simply quiescent, and my view (and that of almost all the designers I have spoken to) is that running it at any higher voltage is simply asking for trouble. This is a particular concern in the design of mixing consoles, which may contain thousands of opamps –anything that impairs their reliability is going to cause a lot of trouble. In any case, moving from ±17 V rails to ±18 V rails only gives 0.5 dB more headroom. Stretching things to ±20 V would give 1.4 dB more than ±17 V, and running on the ragged edge at ±22 V would yield a more significant 2.2 dB more than ±17 V, but you really wouldn’t want to do it. Pushing the envelope like this is also going to cause difficulties if you want to run opamps with maximum supply ratings of ±18 V from the same power supply.
We will therefore concentrate here on ±17 V supplies for opamps, dealing first with what might be called “small power supplies”, i.e. those that can be conveniently built with TO-220 regulators. This usually means an output current capability that does not exceed 1.5 amps, which is plenty for even complicated preamplifiers, electronic crossovers, etc., but will only run a rather small mixing console; the needs of large consoles are dealt with later in this chapter.
An important question is: how low does the noise and ripple on the supply output rails need to be? Opamps in general have very good power supply rejection ratios (PSRR) and some manufacturer’s specs are given in Table 25.1
|Opamp type||PSRR minimum dB||PSRR typical dB|
|+5 V||Housekeeping microcontroller|
|+24 V||LED bar–graph metering systems, discrete audio circuitry, relays|
|+48 V||Microphone phantom power|
The PSRR performance is actually rather more complex than the bare figures given in the table imply; PSRR is typically frequency dependent (deteriorating as frequency rises) and different for the +V and –V supply pins. It is, however, rarely necessary to get involved in this degree of detail. Fortunately, even the cheapest IC regulators (such as the venerable 78xx/79xx series) have low enough noise and ripple outputs that opamp PSRR performance is rarely an issue.
There is, however, another point to ponder; if you have a number of electrolytic-sized decoupling capacitors between rail and ground, enough noise and ripple can be coupled into the non-zero ground resistance to degrade the noise floor. Intelligent placing of the decouplers can help –putting them near where the ground and supply rails come onto the PCB means that ripple will go straight back to the power supply without flowing through the ground tracks on the rest of the PCB. This is of limited effectiveness if you have a number of PCBs connected to the same IDC cable, as in many small mixing desks, and in such cases low-ripple power supplies may be essential.
Apart from the opamp supply rails, audio electronics may require additional supplies, as shown in Table 25.2
It is often convenient to power relays from a +9 V unregulated supply that also feeds the +5 V microcontroller regulator; see later in this chapter. The use of +24 V to power LED metering systems is dealt with in Chapter 23 on metering, and +48 V phantom supplies are examined at the end of this chapter.
Making a straightforward ±15 V 1 amp supply for an opamp-based system is very simple and has been ever since the LM7815/7915 IC regulators were introduced (which was a long time ago). They are robust and inexpensive parts with both overcurrent and over-temperature protection and give low enough output noise for most purposes. We will look quickly at the basic circuit, because it brings out a few design points which apply equally to more complex variations on the theme. Figure 25.1 shows the schematic, with typical component values; a centre-tapped transformer, a bridge rectifier, and two reservoir capacitors C1, C2 provide the unregulated rails that feed the IC regulators. The secondary fuses must be of the slow-blow type. The small capacitors C7–C9 across the input to the bridge reduce RF emissions from the rectifier diodes; they are shown as X-cap types not because they have to withstand 230 Vrms but to underline the need for them to be rated to withstand continuous AC stress. The capacitors C3, C4 are to ensure HF stability of the regulators, which prefer a low AC impedance at their input pins, but these are only required if the reservoir capacitors are not adjacent to the regulators, i.e. more than 10 cm away. C5, C6 are not required for regulator stability with the 78/79 series –they are there simply to reduce the supply output impedance at high audio frequencies.
There are really only two electrical design decisions to be made: the AC voltage of the transformer secondary and the size of the reservoir capacitors. As to the first, you must make sure that the unregulated supply is high enough to prevent the rails dropping out (i.e. letting hum through) when a low mains voltage is encountered but not so high that either the maximum input voltage of the regulator is exceeded, or it suffers excessive heat dissipation. How low a mains voltage it is prudent to cater for depends somewhat on where you think your equipment is going to be used, as some parts of the world are more subject to brown-outs than others. You must consider both the minimum voltage drop across the regulators (typically 2 V) and the ripple amplitude on the reservoirs, as it is in the ripple troughs that the regulator will first “drop out” and let through unpleasantness at 100 Hz.
In general, the RMS value of the transformer secondary will be roughly equal to the DC output voltage.
The size of reservoir capacitor required depends on the amount of current that will be drawn from the supply. The peak-to-peak ripple amplitude is normally in the region of 1 to 2 volts; more ripple than this reduces efficiency, as the unregulated voltage has to be increased to allow for unduly low ripple troughs, and less ripple is usually unnecessary and gives excessive reservoir capacitor size and cost. The amount of ripple can be estimated with adequate accuracy by using Equation 25.1,
- Where Vpk-pk is the peak-to-peak ripple voltage on the reservoir capacitor, I is the maximum current drawn from that supply rail in amps, Δ t is the length of the capacitor discharge time, taken as 7 milliseconds, C is the size of the reservoir capacitor in microfarads, and the “1000” factor simply gets the decimal point in the right place.
Note that the discharge time is strictly a rough estimate and assumes that the reservoir is being charged via the bridge for 3 msec and then discharged by the load for 7 msec. Rough estimate it may be, but I have always found it works very well.
The regulators must be given adequate heat sinking. The maximum voltage drop across each regulator (assuming 10% high mains) is multiplied by the maximum output current to get the regulator dissipation in watts, and a heat sink selected with a suitable thermal resistance to ambient (in °C per watt) to ensure that the regulator package temperature does not exceed, say, 90°C. Remember to include the temperature drop across the thermal washer between regulator and heat sink.
Under some circumstances, it is wise to add protective diodes to the regulator circuitry, as shown in Figure 25.2. The diodes D1, D3 across the regulators are reverse biased in normal operation, but if the power supply is driving a load with a large amount of capacitance, it is possible for the output to remain higher in voltage than the regulator input as the reservoir voltage decays. D1, D3 prevent this effect from putting a reverse voltage across the regulators. Such diodes are not usually required with normal opamp circuitry, as the amount of rail decoupling, shown as C7, C8 in Figure 25.2, is usually modest.
The shunt protection diodes D2, D4 are once again reverse-biased in normal operation. D2 prevents the +15 V supply rail from being dragged below 0 V if the -15 V rail starts up slightly faster, and likewise D4 protects the -15 V regulator from having its output pulled above 0 V. This can be an important issue if rail-to-rail decoupling such as C9 is in use; such decoupling can be useful because it establishes a low AC impedance across the supply rails without coupling supply rail noise into the ground, as C7, C8 are prone to do. However, it also makes a low-impedance connection between the two regulators. D2, D4 will prevent damage in this case but leave the power supply vulnerable to start-up problems; if its output is being pulled down by the -15 V regulator, the +15 V regulator may refuse to start. This is actually a very dangerous situation, because it is quite easy to come up with a circuit in which start-up will only fail 1 time in 20 or more, the incidence being apparently completely random but presumably controlled by the exact point in the AC mains cycle where the supply is switched on, and other variables such as temperature, the residual charge left on the reservoir capacitors, and the phase of the moon. If even one start-up failure event is overlooked or dismissed as unimportant, there is likely to be serious grief further down the line. Every power supply start-up failure must be taken seriously.
There are 15 V IC regulators (7815, 7915), and there are 18 V IC regulators (7818, 7918), but there are no 17 V IC regulators. This problem can be effectively solved by using 15 V regulators and adding 2 volts to their output by manipulating the voltage at the REF pin. The simplest way to do this is with a pair of resistors that divide down the regulated output voltage and apply it to the REF pin as shown in Figure 25.3a. (The transformer and AC input components have been omitted in this and the following diagrams, except where they differ from those shown in Figure 25.1.) Since the regulator maintains 15 V between the OUT and REF pin, with suitable resistor values, the actual output with respect to 0 V is 17 V. Using E24 resistors, R1 = 1 kΩ and R2 = 150 Ω gives a nominal output of +17.25 V, which is near enough for our purposes.
The snag with this arrangement is that the quiescent current that flows out of the REF pin to ground is not well controlled; it can vary between 5 and 8 mA, depending on both the input voltage and the device temperature. This means that R1 and R2 have to be fairly low in value so that this variable current does not cause excessive variation of the output voltage, and therefore power is wasted.
If a transistor is added to the circuit as in Figure 25.3b, then the impedance seen by the REF pin is much lower. This means that the values of R1 and R2 can be increased by an order of magnitude, reducing the waste of regulator output current and reducing the heat liberated. This sort of manoeuvre is also very useful if you find that you have a hundred thousand 15 V regulators in store, but what you actually need for the next project is an 18 V regulator, of which you have none.
What about the output ripple with this approach? I have just measured a power supply using the exact circuit of Figure 25.3b, with 2200 μF reservoirs, and I found -79 dBu (87 μVrms) on the +17 V output rail, and -74 dBu (155 μVrms) on the 17 V rail, which is satisfyingly low for inexpensive regulators and should be adequate for almost all purposes; note that these figures include regulator noise as well as ripple. The load current was 110 mA. If you are plagued by ripple troubles, the usual reason is a rail decoupling capacitor that is belying its name by coupling rail ripple into a sensitive part of the ground system, and the cure is to correct the grounding rather than design an expensive ultra-low-ripple PSU. Note that doubling the reservoir capacitance to 4400 μF only improved the figures to -80 dBu and -76 dBu, respectively; just increasing reservoir size is not a cost-effective way to reduce the output ripple.
It is of course also possible to make a ±17 V supply by using variable output voltage IC regulators such as the LM317/337. These maintain a small voltage (usually 1.2 V) between the OUTPUT and ADJ (shown in figures as GND) pins and are used with a resistor divider to set the output voltage. The quiescent current flowing out of the ADJ pin is a couple of orders of magnitude lower than for the 78/79 series, at around 55 μA, and so a simple resistor divider gives adequate accuracy of the output voltage, and transistors are no longer needed to absorb the quiescent current. A disadvantage is that this more sophisticated kind of regulator is somewhat more expensive than the 78/79 series; at the time of writing they cost something like 50% more. The 78/79 series with transistor voltage-setting remains the most cost-effective way to make a non-standard-voltage power supply at present.
It is clear from Figure 25.4 that the 1.2 V reference voltage between ADJ and out is amplified by many times in the process of making a 17 V or 18 V supply; this increases not only output ripple but also output noise as the noise from the internal reference is being amplified. The noise and ripple can be considerably reduced by putting a capacitor C7 between the ADJ pin and ground. This makes a dramatic difference; in a test PSU with a 650 mA load, the output noise and ripple was reduced from -63 dBu (worse than 78xx series) to -86 dBu (better than 78xx series), and so such a capacitor is usually fitted as standard. If it is fitted, it is then essential to add a protective diode D1 to discharge C7, C8 safely if the output is short-circuited, as shown in Figure 25.5.
The ripple performance of the aforementioned test PSU, with a 4700 μF reservoir capacitor and a 650 mA load, is summarised for both types of regulator in Table 25.3. Note that the exact ripple figures are subject to some variation between regulator specimens. Supply-rail noise and ripple is often quoted in microvolts; see the last column of Table 25.3.
Table 25.3 shows that the best noise and ripple performance that can be expected from a simple LM317 regulator circuit is about -86 dBu (39 μVrms), and this still contains a substantial ripple component. The reservoir capacitors are already quite large at 4700 μF, so what is to be done if lower ripple levels are needed? The options are:
|7815 + transistor dBu||LM317 dBu||LM317 μV|
|No C on LM317 ADJ pin||–73 dBu (all ripple)||–63 dBu (ripple and noise)||549 μV|
|47 μF on LM317 ADJ pin||–73 dBu (all ripple)||–86 dBu (ripple and noise)||39 μV|
|Input filter 2.2Ω and 2200 μF||–78 dBu (ripple and noise)||–89 dBu (mostly noise)||27 μV|
|Input filter 2.2Ω and 4400 μF||–79 dBu (mostly noise)||–90 dBu (all noise)||24 μV|
- Look for a higher-performance IC regulator. They will cost more, and there are likely to be issues with single sourcing.
- Design your own high-performance regulator using discrete transistors or opamps. This is not a straightforward business if all the protection that IC regulators have is to be included. There can also be distressing issues with HF stability.
- Add an RC input filter between the reservoir capacitor and the regulator. This is simple and pretty much bulletproof and preserves all the protection features of the IC regulator, though the extra components are a bit bulky and not that cheap. There is some loss of efficiency due to the voltage drop across the series resistor; this has to be kept low and the capacitance large.
The lower two rows of Table 25.3 show what happens. In the first case, the filter values were 2.2 Ω and 2200 μF. This has a -3 dB frequency of 33 Hz and attenuates the 100 Hz ripple component by 10 dB. This has a fairly dramatic effect on the output ripple, but the dB figures do not change that much, as the input filter does not affect the noise generated inside the regulator. Increasing the filter capacitance to 4400 μF sinks the ripple below the noise level for both types of regulator.
It is extremely convenient to use third-party “wall-wart” power supplies for small pieces of equipment, as they come with all the safety and EMC approvals already done for you, though admittedly they do not look appropriate with high-end equipment.
The problem is that the vast majority of these supplies give a single AC voltage on a two-pole connector, so a little thought is required to derive two supply rails. Figure 25.6 shows how it is
done in a ±18 V power supply; note that these voltages are suitable only for a system that uses 5532s throughout. Two voltage-doublers of opposite polarity are used to generate the two unregulated voltages. When the incoming voltage goes negative, D3 conducts and the positive end of C1 takes up approximately 0 V. When the incoming voltage swings positive, D1 conducts instead and the charge on C1 is transferred to C3. Thus the whole peak-to-peak voltage of the AC supply appears across reservoir capacitor C3. In the same way, the peak-to-peak voltage, but with the opposite polarity, appears across reservoir C4.
Since voltage-doublers use half-wave rectification, they are not suitable for high current supplies. When choosing the value of the reservoir capacitor values, bear in mind that the discharge time in Equation 25.1 must be changed from 7 msec to 17 msec. The input capacitors C1, C2 should be the same size as the reservoirs.
One of the main reasons for using discrete audio electronics is the possibility of handling larger signals than can be coped with by opamps running off ±17 V rails. The use of ±24 V rails allows a 3 dB increase in headroom, which is probably about the minimum that justifies the extra complications of discrete circuitry. A ±24 V supply can be easily implemented with 7824/7924 IC regulators.
A slightly different approach was used in my first published preamplifier design.  This preamp in fact used two LM7824 +24 V regulators connected as shown in Figure 25.7, because at the time, the LM7924 –24 V regulator had not yet reached the market. The use of a second positive regulator to produce the negative output rail looks a little strange at first sight, but I can promise you it works. It can be very useful in the sort of situation described earlier; you have a hundred thousand +15 V regulators in store but no -15 V regulators … I’m sure you see the point.
Note that this configuration requires two separate transformer windings; it cannot be used with a centre-tapped secondary.
So what if you need more than 1 amp of current? This will certainly be the case for all but the smallest mixing consoles. There are of course IC regulators with a greater current capability than 1 amp. The LM338K variable-voltage 5 amp regulators come in a TO-3 package –which, it has to be said, is not the easiest format to mount. They were once widely used in mixing console power supplies but have been superseded by more modern devices such as the Linear Tech LT1083CK, which is a 7.5 amp variable-voltage regulator that works very nicely and can be obtained in a TO-3P package. I have used lots of them in mixing console power supplies. An example of their use, incorporating a mutual-shutdown facility, is shown in Figure 25.8.
There is no negative version of the LT1083CK, so a +/- supply has to be made with two separate secondary windings, just as in Figure 25.7. When using any high-current regulator, it pays to be rather cautious about the maximum rated current. Trying to use them too close to the maximum can cause start-up problems.
It is an awkward quirk of 5532 opamps that if one supply rail is lost and collapses to 0 V, while the other rail remains at the normal voltage, they can, under some circumstances, get into an anomalous mode of operation that draws large supply currents and ultimately destroys the opamp by over-heating. To prevent damage from this cause, which could be devastating to a large mixing console, the opamp supplies are very often fitted with a mutual shutdown system. Mutual shutdown ensures that if one supply rail collapses because of overcurrent, over-temperature, or any other cause, the other rail will be promptly switched off. The extra circuitry required to implement this is shown in Figure 25.8, which is an example of a high-current supply using 7.5 amp regulators.
The extra circuitry to implement mutual-shutdown in Figure 25.8 is very simple; R5, D3, R6, and Q1 and Q2. Because R5 is equal to R6, D3 normally sits at around 0 V in normal operation. If the +17 V rail collapses, Q2 is turned on by R6, and the REF pin of U2 is pulled down to the bottom rail, reducing the output to the reference voltage (1.25 V). This is not completely off, but it is low enough to prevent any damage to opamps.
If the -17 V rail collapses, Q1 is turned on by R5, pulling down the REF pin of U1 in the same way. Q1 and Q2 do not operate exactly symmetrically, but it is close enough for our purposes.
Note that this circuit can only be used with variable-output voltage regulators, because it relies on their low reference voltages.
By “very large”, I mean too big to be implemented with IC regulators –say, 7 amps and above. This presents a difficult problem, to which there are several possible answers:
- Split up the system supply rails so that several IC regulators can be used. This is, in my view, the best approach. The amount of design work is relatively small; in particular, the short-circuit protection has been done for you.
- Use power transistors as series-pass elements, controlled by opamps. This can be a surprisingly tricky technology. The feedback system has to be reliably stable, and the short-circuit protection has to be foolproof. Designing the latter is not too simple.
- Switch-mode power supplies. Those found in PCs seem to be a mature technology and are very reliable. Custom designs, of the sort required for this application, are in my experience another matter. I have seen them explode. There is also the issue of RF emissions.
The first two methods obviously involve increased heat dissipation in proportion to their output current. This usually means that fan cooling is required to keep the heat sinks down to a reasonable size, which is fine for PA work but not welcome in a studio control room. There are no great technical difficulties to powering even a large console over a 20-metre cable (this is the recommended maximum for Neve consoles) so long as remote sensing is used to compensate for voltage drops. 20 metres is usually long enough to allow the power supply to be placed in another room.
The technology of very large audio supplies is a specialised and complicated business, and it would not be appropriate to dig any further into it here.
It is very often most economical to power relays from an unregulated supply. This is perfectly practical, as relays have a wide operating voltage range. If 9 V relays are used, then the same unregulated supply can feed a +5 V regulator to power a microcontroller, as shown in Figure 25.9.
Hum induced by electrostatic coupling from an unregulated relay supply rail can be sufficient to compromise the noise floor; the likelihood of this depends on the physical layout, but inevitably the signal paths and the relay supply come into proximity at the relay itself. It is therefore necessary to give this rail some degree of smoothing without going to the expense of another regulator and heat sink. (There must be no possibility of coupling between signal ground and relay power ground; these must only join right back at the power supply.) This method of powering relays is more efficient than a regulated rail, as it does not require a voltage drop across a regulator that must be sufficient to prevent drop-out and consequent rail ripple at low mains voltages.
Simple RC smoothing works perfectly well for this purpose. Relays draw relatively high currents, so a low R and a high value C are used to minimise voltage losses in R and changes in the relay supply voltage as different numbers of relays are energised.
The RC smoothing values shown in Figure 25.9 are typical but are likely to need adjustment depending on how many relays are powered and how much current they draw. R1 is low at 2.2 Ω and C2 high at 4700 μF; fortunately, the voltage is low, so C2 need not be physically large.
Making a discrete +48 V regulator with the necessary low amounts of noise and ripple is not too hard, but making one that is reliably short circuit–proof is a little more of a challenge, and by far the easiest way to design a +48 V phantom supply is to use a special high-voltage regulator called the TL783C, as shown in Figure 25.10. This extremely handy device can supply 700 mA, subject to power dissipation constraints.
It is a variable-voltage device, maintaining typically 1.27 V between the OUT and ADJ pins. It combines BJT circuitry with high-voltage MOS devices on the same chip, allowing it to withstand much higher voltages than standard bipolar regulators. Since MOS devices are not subject to secondary breakdown or thermal runaway, the TL783 still gives full overload protection while operating with up to a 125 V voltage drop from the input to the output. The TL783 has current limiting, safe-operating-area (SOA) protection, and thermal shutdown. Even if the ADJ pin is accidentally disconnected, the protection circuitry stays operational. It is a very useful and reliable IC, and I have deployed thousands of them.
As with other variable-voltage regulators, the low voltage maintained between the OUT and ADJ pins needs to be amplified by a considerable ratio to get the desired output voltage, and so the reference voltage tolerances are also amplified. In this case the amplification factor is as high as 37 times, and so a preset is used to adjust the output voltage to exactly +48 V. The filter capacitor C3 is essential for the same reason: without it, the ripple is amplified along with the reference voltage.
The unregulated supply can be derived from a completely separate transformer secondary as in Figure 25.10 or, alternatively, by means of a voltage doubler. The latter is usually more economic, but obviously this depends on the cost of an extra transformer winding versus the cost of the extra capacitor in the doubler.
The arrangement of a voltage-doubler phantom supply is shown in Figure 25.11. Note that the familiar voltage doubler circuit C13, C2, D5, D6 is actually working as a voltage tripler, because it is perched on the unregulated +23 V supply to the +17 V regulator. If it worked as a true voltage doubler, based on the 0 V rail, it would generate insufficient unregulated voltage for the phantom regulator. Because of their inherently half-wave operation and relatively poor regulation, voltage doubler or voltage tripler methods are not suitable for high-current phantom supplies.
 Self, D. R. G. “An Advanced Preamplifier Design” Wireless World, Nov 1976